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Inverting Converter
Design
The Design of Inverting DC to DC Converters
Before reading this page, please read the
introduction.
All of
the circuits in this tutorial can be simulated in
LTspice®. If you are new to
LTspice, please have a
look at my
LTspice Tutorial
Introduction
There are several ways of generating a negative
voltage from a positive one, each with its own
merits and drawbacks. This article will discuss the
two popular architectures: the single inductor
inverter and the Cuk (pronounced Chook)
Converter (named after its inventor, Dr. Slobodan
Ćuk).
As with all dc/dc converters, each one relies on the
flyback properties of an inductor to generate the
negative voltage.
(The Cuk converter is a
superior solution to the single inductor inverter,
but requires 2 inductors which is not desirable in
some cases. As the two circuits have different ways
of operating, each converter is described in detail,
right from the basics, which makes this page rather
long. If you want to skip to the Cuk converter,
please scroll down and start reading from there)
Simple Inverter
A simple, single inductor based inverter is shown in
FIG 1. This circuit converts a 5V input to -5V.
FIG 1
The LTspice circuit of FIG 1 can be downloaded here:
Simple
Inverting dc/dc converter.
The datasheet of the LT3481 can be downloaded here:
LT3481 datasheet.
An internal transistor switches ON connecting Vin to
SW, thus applying the input voltage across the
inductor L1. The current ramps through the inductor
according to the equation
where Vin is the input voltage (and the voltage
across the inductor), L is the inductance value in
Henries and di/dt is the change
in current with time, measured in Amps per second.
In the circuit in FIG 1, the above equation becomes
or 1.063 million amps per second. If the internal
switch switches off after 1us, the current will have
ramped up by 1.063A.
The LTspice simulation shows this current is 1.02
million amps per second. The slight error is due to
the voltage drop across the switch causing a voltage
slightly less than Vin being applied to the
inductor.
When the transistor switches OFF, the inductor tries
to maintain its current flow. It does this by
generating a voltage across its terminals very
similar to a battery, where the current flows from
the negative terminal, through the battery, to the
positive terminal. Since the right hand side of the
inductor is clamped to 0V, the left hand side of the
inductor flies negative. The Schottky diode, D1,
conducts and clamps the left hand side of the
inductor to about 0.3V below Vout and a current
circulates clockwise down through the capacitor, up
the diode and from left to right through the
inductor, thus charging the capacitor. Since the
upper terminal of the capacitor is at 0V, the lower
terminal charges negatively and a negative voltage
appears at the OUT terminal. The internal switch
then switches ON again and the process starts over.
The inductor discharges according to the equation
(ignoring the diode drop).
Thus during discharge the change in current with
time is also 1.063 million amps per second and this
can be seen in LTspice.
It is interesting to note that the value of di/dt
is determined ONLY by the inductance value and
the voltage across the inductor. The controller
IC has nothing to do with setting the inductor ramp
current.
The voltage across capacitor C1 is monitored by
resistors R1 and R2 and when the junction of R1 and
R2 reaches 1.265V (see LT3481 datasheet), the
switching stops. Thus by setting R1 and R2 we can
determine the final (negative) output voltage at
OUT.
It is worth noting that the circuit in FIG 1 is very
similar to a buck converter. Indeed the LT3481 is
advertised as a buck converter and not an inverter.
The only difference is that the node that was the
output in the buck converter is now shorted to
ground and the node that was 0V (including the
ground pin of the controller) is now the negative
output. Thus the ground pin of the LT3481 moves down
in voltage as the capacitor charges, but the input
voltage ground and the output voltage ground are the
same node, so it is OK to short them together
without things going bang. The input voltage is
still referenced to 0V though.
A standard buck converter can be used as an inverter
because the phasing of the feedback pin does not
change. As the output capacitor charges negatively,
the ground reference of the controller (the ground
pin) is pulled negative, thus the voltage on all the
other pins rise with respect to the ground pin. Thus
the junction of R1 and R2 rises as the output
capacitor charges – the same as a buck converter.
Compare this with the circuit of FIG 2. This is an
alternative solution to the single inductor
inverting dc/dc converter. Architecturally this is
nearly identical to the circuit in FIG 1.
FIG 2
The input voltage is applied one end of the inductor
(with the other end at ground) and the diode
conducts on the inductor discharge cycle so the
voltage at the OUT pin ramps to a negative voltage.
Even the input and output share the same ground.
However, here the feedback resistors, R1 and R2, are
referenced to the REF pin (which is usually a
positive voltage of approximately 1.2V). With 0V on
the output, the junction of R1 and R2 is at a
positive voltage and ramps negatively (so a standard
buck converter cannot be used) as the OUT pin ramps
to a negative voltage. R1 and R2 are scaled such
that the junction of the resistors is at 0V when the
output voltage reaches regulation.
The disadvantage of this architecture is that the
REF pin has to source current into the feedback
network and this might affect its accuracy, hence
the accuracy of the output voltage. The accuracy of
the output voltage is also dependent on the accuracy
of the feedback resistors. The accuracy of the
circuit in FIG 1 is only dependent on the accuracy
of the feedback resistors. However, with the circuit
in FIG 1, the controller is exposed to a supply
voltage equal to the input voltage plus the
magnitude of the output voltage, which can be quite
high, but this is not normally a problem.
The input current (in green) and inductor current
(in blue) for the circuit in FIG 3 are shown below.
FIG 3
It is useful to determine the duty cycle of the
converter. This is the ratio of the ON time of the
switch to the total switching period.
The inductor charges according to
and discharges according to
(ignoring the diode drop). We are also considering
the magnitude of Vout to make the equations easier.
Thus, if the change in charge current is equal to
the change in discharge current then
so
where dt1 is the charge time and dt2
is the discharge time of the inductor.
If we define the total switching period (dt1+dt2)
as T then the duty cycle (DC) is
Therefore
so
becomes
and from here it can be found that
where Vout is the magnitude of Vout.
Again, the duty cycle is set by the input and output
voltages only. The inductor value does not
feature in setting the duty cycle, nor does the
controller IC. This is true as long as the
current in the inductor does not fall to zero. This
is called Continuous Conduction Mode (CCM). If the
inductor current falls to zero, the duty cycle
equation above does not hold and the controller
enters Discontinuous Conduction Mode (DCM).
In CCM, if the load current increases, the duty
cycle remains unchanged (in steady state). The
circuit reacts to the increase in load current by
keeping the duty cycle constant, but the midpoint of
the inductor current (its dc offset) increases. The
switching frequency and the amplitude of the
inductor ripple current remain unchanged. In FIG 3,
the midpoint of the inductor current is
approximately 1.1A and the ripple amplitude is
700mA. If the load increases the midpoint of the
current will increase, but the inductor ripple
current will still be 700mA and the duty cycle will
remain unchanged.
Now, the LT3481 is a buck converter and we have
already stated that the circuit in FIG 1 is similar
to that of a buck converter. Indeed it has the
inductor/diode/capacitor configuration of a buck
converter. With a buck converter the average
inductor current (equal to the mid point of the
inductor current) is also equal to the output
current. Since FIG 1 is so similar to a buck
converter, it would be convenient to assume that the
average inductor current is also equal to the output
current for the inverting configuration. However FIG
3 shows the average inductor current as
approximately 1.1A when we know that the output
current of FIG 1 is 500mA. The inverting
configuration has a higher average inductor current
because the inductor is actually disconnected from
the output while it is charging. Consider a buck
converter architecture shown in FIG 4. During the
inductor charge phase, current flows through Q1,
through the inductor and into the output capacitor.
During the discharge phase, current flows up through
Q2, through the inductor and into the output
capacitor. On both the charge and discharge phase,
the capacitor is always receiving current.
FIG 4
With the inverting configuration, shown in FIG 5, on
the inductor charge phase, the current flows out of
the SW pin through the inductor and down to ground.
The output only receives current during the
discharge phase. The shorter the duty cycle, the
more time the discharge cycle can dump current into
the load. Therefore, the average current in the
inductor can be represented by
With a load current of 500mA (in FIG 1), and a duty
cycle of 50%, the average inductor current will be
1A (assuming no losses).
FIG 5
The single inductor inverter is simple, but due to
the higher inductor current and sharp changes in
current (see the green waveform in FIG 3), it does
not present an elegant way of generating a negative
voltage. A more suitable solution (the Cuk
converter) will be discussed later.
For completion, below is a design example of a
single inductor inverter.
Single Inductor Inverter Design Procedure
Our
design brief is to design an inverting controller to
convert 12V to -5V at 1A with a switching frequency
of 400kHz.
An outline schematic is shown in FIG 6.
FIG 6
The LTC3854 is a standard buck converter, but in FIG
6 is wired as an inverter to give -5V/1A at the OUT
terminal.
Inductor Choice
With an input voltage of 12V and an output voltage
of -5V, the duty cycle is represented by
The LTC3854 switches at a frequency of 400kHz, so
the ON time of the top MOSFET is 29% of 2.5us, or
725ns. A check of the datasheet shows that the
minimum ON time of the LTC3854 is 75ns, so we are
well within spec.
From
the average inductor current will be 1.41A
The optimal ripple current of the inductor is 40% of
the output current. This is a good rule of thumb for
most dc/dc converters and represents a trade off
between small inductor size and low switching
losses. Therefore our design should have an inductor
ripple current of 0.56A.
From the equation
during the charge phase the voltage across the
inductor is equal to the input voltage, the value of
di is 0.56A as calculated above and dt
is 725ns. This means our ideal inductor value needs
to be 15.53uH. With an average inductor current of
1.41A and a peak to peak inductor current of 0.56A,
implies the peak current is 1.69A (and a trough
current of 1.13A)
Now, if too much current flows in the inductor, the
ferrite that it is wound on saturates with the
effect that its inductance rapidly decreases. From
the equation above, if the inductance decreases the
change in current with time increases, worsening the
effect of the over current, so we must ensure that
the inductor we choose is rated to handle the
current. Thus the saturation rating of the inductor
needs to be in excess of the peak current of 1.69A.
A saturation rating over 2A should suffice.
Wurth have a 15uH inductor with a saturation
current rating of 2.2A
744065150 Datasheet
Rsense Calculation
As the inductor current ramps up it develops a
voltage across the current sense resistor R3. The
top MOSFET switches off when the voltage across the
current sense resistor is 50mV (see LTC3854
datasheet). As stated above, the peak inductor
current should be less than 2.2A, so a current sense
resistor of 25mOhms ensures the peak current will be
less than 2A.
MOSFET Choice - General
In nearly all applications the specification for the
top MOSFET is different from that for the bottom
MOSFET if maximum efficiency is to be achieved.
Both MOSFETs will be exposed to the difference
between the input voltage and the output voltage at
some point during the switching cycle, so must both
have a drain-source breakdown voltage of at least
(Vin + magnitude of Vout). In our design, the input
voltage is 12V and the output voltage is -5V, so the
minimum breakdown voltage should be 17V. A MOSFET
rated with a breakdown voltage of at least 30V
should suffice.
The peak current will occur just as the top MOSFET
switches off and the bottom MOSFET switches on and
the same magnitude of current flows through both
devices. Our current sense resistor sets the peak
current to 2A, so any MOSFET with a peak current
greater than 5A is suitable.
Looking at the block diagram of the LTC3854, we see
that the drive circuitry for the bottom MOSFET is
powered from INTVCC. The minimum voltage
specification on this voltage is 4.8V, so our bottom
MOSFET must have a gate turn on voltage of
significantly less than 4.8V.
However, the drive to the top MOSFET is powered from
INTVCC – 0.3V (the voltage across the flying
capacitor) so the turn on voltage of the top MOSFET
needs to be significantly less than 4.5V.
In either case, a logic level MOSFET, with a turn on
voltage of 1V - 2V is more suitable.
The above parameters represent the bare minimum
characteristics of the MOSFETs. However, to get a
good design, we must ensure that the losses in the
MOSFETs are as low as possible.
MOSFET Choice – Switching and Conduction Losses
The MOSFET switches present 2 losses in the circuit:
switching losses and conduction losses.
The switching losses result from current flowing
through the MOSFET at the same time that a voltage
is across the MOSFET (so power is generated in the
MOSFET), during the turn on and turn off times of
the MOSFET. For a given gate drive coming out of the
controller IC, the lower the Gate-Source capacitance
of the MOSFET, the quicker the MOSFET will turn on.
Thus the Qg specification of the MOSFET is important
and should be as low as possible. The Qg of the
MOSFET will also have an impact on the heat
dissipation of the chip, especially if the input
voltage to the chip is high.
Charge is dictated by the equation:
Charge (Q) = Current (I) x Time (s)
Since Frequency is the inverse of Time, we can write
So we can calculate the current needed to flow into
the chip, just to charge the gate capacitance of the
FET. Since heat is the product of voltage and
current, if the gate charge is high and/or the
switching frequency is high, the heat dissipation in
the chip will be high.
Once the MOSFET has switched on, the MOSFET presents
a small dc resistance between its Drain and Source
terminals. This is the MOSFETs ‘Drain Source ON
resistance’ or RDSON. Again, this needs to be as low
as possible.
Now, MOSFET manufacturers reduce the ON resistance
of the MOSFET by constructing many parallel
conduction paths between the Drain and Source. Thus,
like connecting resistors in parallel, the ON
resistance comes down with more parallel paths.
However, in connecting Drain Source paths in
parallel, a negative effect is that the Gate Source
capacitance (Qg) is also connected in parallel, so a
low ON resistance (and hence low conduction loss)
sometimes implies a high gate source capacitance
(hence high switching loss). Thus the MOSFET that is
chosen should be a compromise between these two
characteristics. In addition, high current MOSFETs
tend to come in much larger packages, so meeting the
ideals of low ON resistance and low Qg might violate
a space requirement spec, so the selection process
has to start over. Engineering, as ever, is a
compromise.
Indeed looking at the selection tables of the MOSFET
manufacturers, it is better to select a MOSFET with
a low ON resistance (less than 10mOhms), then filter
this selection to remove MOSFETs with a Qg of
greater than 10nC, then select a MOSFET from this
list, as long as the Gate turn on voltage, Vds and
Id can be met. Starting by selecting MOSFETs with a
Vds of between 20V and 30V might rule out some
higher voltage FETs that are better suited to lower
voltage designs.
Failing that, download all the results to a
spreadsheet and sort from there. I have never had
much luck with the parametric searches on MOSFET
websites.
Alternatively, download all the MOSFET
characteristics into a spreadsheet, remove the ones
that don't meet the VDS and ID requirements, then
add a column called FOM (Figure of Merit). This
column should contain the value RDSON x QG. Then
sort by this column and pick the FET with the lowest
FOM. This part will be the best compromise between
RDSON and QG and ideal for the top MOSFET.
MOSFET Choice – Top MOSFET
The Duty Cycle governs how long the top MOSFET
switches on for per period of the switching
frequency. We have calculated that the duty cycle is
dictated by the ratio of Vout to (Vin+Vout) (when
operating in continuous conduction mode). So it can
be argued that if the input voltage is high and the
output voltage is low (i.e. a low duty cycle),
conduction losses in the top MOSFET are not
important since the top MOSFET is only ON for a
short amount of time. Therefore for low duty cycle
circuits, a MOSFET with low Qg should be chosen,
almost regardless of RDSON. Although there is no
figure as to what constitutes a low duty cycle, any
circuit with a duty cycle of less than about 15%
warrants having its MOSFET optimised for low Qg with
RDSON being largely unimportant.
That said, our duty cycle is 29%, so unfortunately
we should strive to find a MOSFET with both low Qg
and low ON resistance!
Luckily the LTspice model comes with an extremely
good top MOSFET model, the Renesas RJK0305. This
device has an RDSON of 6.7mOhms and a Qg of 8nC.
RJK0305
Datasheet
MOSFET Choice – Bottom MOSFET
When the top MOSFET switches off, the voltage at the
left hand side of the inductor flies negative, thus
the voltage across the bottom MOSFET is nearly zero
when the bottom MOSFET switches on. Therefore the
switching losses of the bottom MOSFET are
negligible, so we do not have to worry about the Qg
specification of the bottom MOSFET. Only the RDSON
characteristic of the bottom MOSFET is important.
In fact, every MOSFET has a ‘body diode’. This is a
diode inherent in the structure of the MOSFET and in
an N channel FET, its anode is connected to the
source and the cathode is connected to the Drain.
When the inductor voltage flies negative, it is the
body diode that conducts first before the gate drive
to the MOSFET activates the Drain-Source channel.
FIG 7 shows a simulation of the switch node just as
the bottom MOSFET is switching on.
FIG 7
We can see the switch node (V(sw)) falling to a
voltage below V(out) – in green - well before the
drive to the bottom MOSFET gate starts to rise (in
red). This is indicative of the body diode starting
to conduct and indeed the negative voltage is
approximately -0.6V. When the body diode conducts,
it stores charge in the MOSFET that has to be
removed before the MOSFET can fully turn on, so body
diode conduction can affect the efficiency of the
converter.
If optimum efficiency is desired, it is wise to
place a Schottky diode across the bottom MOSFET, so
the Schottky diode can conduct the inductor flyback
voltage and not the body diode. The resulting
increase in efficiency can be as much as 3%. The
Schottky diode will conduct the peak current flowing
through the inductor, but this current will only
flow for a short period of time (until the bottom
MOSFET switches on). Therefore, the current rating
of the diode can be a lot less than peak inductor
current. An MBRS340 has a reverse voltage rating of
40V, but a non repetitive peak forward current of
40A.
MBRS340
Datasheet
For the bottom MOSFET, the Renesas RJK0301 has
2.3mOhms RDSON and a Qg of 32nC.
RJK0301 Datasheet
Output Capacitor Choice
During the charging cycle of the inductor, the
output capacitor has no current flowing into it,
similar to a boost converter. Therefore the load
current comes purely from the output capacitor. When
the inductor discharges, the output capacitor it is
subjected to an inrush current. If the capacitor has
any ESR (effective series resistance) this will
develop a ripple voltage on the output capacitor.
Therefore the output ripple is made up of 2
components: the ripple caused by the output
capacitor discharging when the inductor is being
charged and the ripple caused by the inrush current
from the inductor into the ESR of the output
capacitor.
The ripple caused by the discharge of the output
capacitor while the inductor is charging is dictated
by
where i is the load current in Amps, C
is the output capacitance in Farads and dv/dt
is the change in output voltage with time.
Earlier we calculated that the MOSFET switches on
for a period of 725ns. If we require a discharge
ripple of 0.5% (25mV) with a load current of 1A,
this implies we need a capacitance of
or 29uF.
Note that when the inductor is charging, there is
zero current flowing in the output capacitor. When
the top MOSFET switches off, the bottom MOSFET
current (and hence the capacitor charge current)
jumps from 0A to the peak inductor current, so it is
the peak inductor current, not the ripple current
amplitude that determines this component of the
output ripple (compare this with the ripple in a
buck converter that is determined by the ripple
current amplitude, not the peak inductor current).
The ripple caused by the ESR is a product of the
peak inductor current and the ESR. In our example
the peak current is 1.69A and the ESR is of a
typical tantalum capacitor is 70m Ohms, giving a
ripple of 118mV.
At this point is it worth trading off ESR ripple for
discharge ripple and repeating the above two
calculations on several combinations of output
capacitor to see the effect on the ripple. We see
that it is relatively easy to meet the spec of
discharge ripple since our output capacitor is only
29uF. However if we are to achieve an overall output
ripple of 1% (50mV) we are going to need lots of
capacitors in parallel to meet the ESR required to
keep the ESR ripple low.
Repeating the above calculations several times, it
would appear that three 22uF capacitors in parallel
will meet our overall ripple spec. 66uF will give a
discharge ripple of 11mV and if each capacitor has
an ESR of 70mOhms, the ESR ripple will be 39mV.
Other Points
The feedback resistor values can be calculated using
this spreadsheet:
Feedback Resistor Calculator
The final circuit is shown in FIG 8.
FIG 8
The LTspice circuit in FIG 8 can be downloaded here:
Single Inductor
Inverting dc/dc Converter
The Cuk Converter
In FIG 1 it can be seen that although the inductor
current has a smooth charge/discharge, the input
current (measured by holding down the ALT key and
probing the input current into the controller) has
sharp rise and fall times. This can lead to
interference being generated in the circuitry
supplying the input current. This is where the Cuk
Converter provides a more suitable alternative.
The Cuk converter has an inductor on the input and
the output, so both input and output currents have
no sharp changes in current.
FIG 9 shows a Cuk Converter.
FIG 9
The LTspice Cuk Converter circuit above can be
downloaded here: Cuk
Converter.
It can be seen that the circuit about has the input
configuration of a boost converter and the output
configuration of a buck converter. However, unlike a
boost or buck converter, the controller in a Cuk
converter needs to be able to respond to a negative
feedback voltage.
When MOSFET Q1 switches on, the right hand side of
inductor L1 is shorted to ground. The current in the
inductor ramps according to the equation
where V is the voltage across the inductor (in this
case it is equal to the input voltage), L is the
inductor value and di/dt is the change in
inductor current with time. Thus with a fixed
voltage across the inductor and a fixed inductor
value, the change in current with time is constant.
Thus in FIG 9 the change in current with time can be
represented by
Or 700,000 Amps per second.
When the MOSFET switches off, the inductor tries to
maintain its current flow. It does this by creating
a voltage across it where the right hand side tries
to fly positive (to push current out of the right
hand end) and the left hand side flies negative.
Since the left hand side of the inductor is clamped
to the input voltage, the right hand side of the
inductor flies positive to a voltage above Vin in
order to maintain current flow. The energy from the
inductor flows into capacitor C5 charging it with a
positive voltage (which is higher than Vin). The
right hand side of C5 is clamped to +0.3V by diode
D1, but for the sake of convenience we will ignore
this voltage drop and assume the right hand side of
the capacitor is clamped to 0V. We will work out
later exactly what voltage C5 charges to, but for
the moment it is sufficient to assume it charges to
a voltage higher than Vin. We will call this voltage
Vcap.
Since the voltage Vcap is higher than Vin, the
voltage across the inductor now has the opposite
polarity to before. The inductor discharges
according to the equation
where V is the voltage across the inductor, thus
It is interesting to note that the value of di/dt
is determined ONLY by the inductance value and
the voltage across the inductor. The controller
IC has nothing to do with setting the inductor ramp
current.
When the MOSFET switches on again the voltage on the
drain of the MOSFET goes from Vcap to 0V. Since the
voltage across a capacitor cannot change
instantaneously, an equal negative going voltage
appears on the anode of diode D1 so this node
transitions from 0V to –Vcap. We now have a negative
amplitude square wave voltage (at the right hand
node of C5) being applied to an LC filter (L2 and
C1). The LC filter averages out this square wave to
produce a flat DC voltage whose amplitude is
somewhere between 0V and –Vcap. This amplitude is
dictated by the duty cycle of the square wave.
We are now going to calculate the duty cycle (the
ratio of the ON time of the MOSFET to the total
switching period) and the voltage (Vcap) on the
coupling capacitor C5.
The inductor charge and discharge currents are equal
when the circuit is in steady state. Therefore
where dt1 is the ON time of the MOSFET
and dt2 is the OFF time of the MOSFET.
Dividing both sides by (dt1+dt2)
gives
If the Duty Cycle (DC) can be represented by
then
so
To determine the duty cycle in terms of the input
and output voltages, consider FIG 10
FIG 10
Here we can see the Drain voltage going from 0V to
Vcap (as yet uncalculated) and the ac coupled drain
voltage on the anode of the diode. The capacitor has
removed the dc offset and the diode has clamped the
positive excursions to roughly 0V.
Now, when the circuit is regulating there will be a
flat negative dc voltage on the output. Thus, when
V(diode) is at 0V there will be a positive voltage
from V(diode) to V(out) and the inductor current in
L2 will ramp in a positive direction. When V(diode)
is negative there will be a negative voltage from
V(diode) to V(out) so the inductor current will ramp
to a more negative value.
In steady state, when the MOSFET switches ON V(diode)
is at –Vc and the voltage across inductor L2 is
(-Vout-(-Vcap)), thus the change in current is
represented by
When the MOSFET switches OFF, the voltage across L2
is (0-(-Vout)), so the change in current is
represented by
Equating the values of di gives
Dividing both sides by (dt1 + dt2)
gives
where DC is the duty cycle as defined above.
Thus
From before we know that
So
So
Vout is the magnitude of the output voltage.
This is because in the above derivation, we have
ignored the slope of di – it is positive in
L1 when negative in L2, so cannot strictly equate
the 2 statements for DC without considering this.
The result of knowing Vcap is that we now know that
the Drain of the MOSFET is exposed to a voltage
equal to (Vin + |Vout|) and has to be sized
accordingly (as does the capacitor’s working
voltage).
Knowing that
and
We can work out the Duty Cycle in terms of Vout and
Vin. Thus
Again, Vout is the magnitude of the output voltage.
The duty cycle is set by the input and output
voltages only. The inductor value does not
feature in setting the duty cycle, nor does the
controller IC.
The above is true as long as the current in the
inductor does not fall to zero. This is called
Continuous Conduction Mode (CCM). If the inductor
current falls to zero, the duty cycle equation above
does not hold and the controller enters
Discontinuous Conduction Mode (DCM).
In CCM, if the load current increases, the duty
cycle remains unchanged (in steady state). The
circuit reacts to the increase in load current by
keeping the duty cycle constant, but the midpoint of
the inductor current (its dc offset) increases. The
switching frequency and the amplitude of the
inductor ripple current remain unchanged.
Cuk Converter Design Procedure
Our design brief is to design a Cuk Converter with
an input voltage of 10V and an output voltage of 5V
that can support a load of 1A. The switching
frequency should be 300kHz.
FIG 11 shows a Cuk Converter based on the LT3757.
FIG 11
With a 10V input and a 5V output, we can calculate
the duty cycle (DC) as being
With a switching frequency of 300kHz, this
represents a period of 3.33us, so with a duty cycle
of 33% the MOSFET ON time is 0.33 x 3.33us = 1.11us.
The minimum ON time of the LT3757 is 220ns, so this
is well within spec.
Inductor Choice
It is good design practice to keep the ripple
current in the inductor at 40% of the total current.
This is a good trade off between small inductor size
and low switching losses. The inductor on the output
of a Cuk Converter is configured identically to that
of a buck converter. With the buck converter, the
average inductor current is equal to the output
current. On the input, the Cuk Converter has an
inductor configured identically to that of a boost
converter and the average inductor current in a
boost converter is equal to the average input
current.
With an output voltage of 5V and a load of 1A, this
represents an output power of 5W. Allowing for an
efficiency of 85% for the converter, this means our
input power has to be 5.88W. With an input voltage
of 10V, this represents an average input current of
588mA.
If the input inductor current ripple is 40%, then
the peak inductor current is 588mA x 1.2, or 706mA
and the trough inductor current is 588mA x 0.8, or
470mA.
The change in current is therefore 236mA.
From
when the MOSFET switches ON, the voltage across the
input inductor is 10V, we have calculated that the
ON time of the MOSFET is 1.11us, so we can work out
the value of input inductance needed to get a change
in current of 236mA. Thus
To calculate the output inductor value, we go
through the same procedure.
We know that
and we know the voltage on the anode of D1 in FIG 11
is a square wave with amplitude of Vcap, we know
that the output inductor has a voltage across it of
Vcap – Vout (=Vin) when the MOSFET is ON, so for the
same ON time our output inductor should be the same
value as the input inductor for the same change in
current. The purists would argue that since our
output current is different to the input current
then keeping both inductor values the same will
result in a different ripple percentage in the
output inductor, so the output inductor could be
sized differently to reflect this, but the resulting
change in circuit performance is minimal for most
applications.
However, it should be noted that the current in the
output inductor is considerably higher in this case
(since we are stepping down the voltage, so
stepping up the current). Therefore, if the
average output inductor current is equal to the
output current and we have a ripple current of
236mA, our peak output inductor current will be (1A
+ 118mA) = 1.12A.
So our input inductor needs to have a saturation
current rating of at least 706mA and our output
inductor needs to have a saturation current rating
of at least 1.12A. It is convenient to select 2
identical inductors (for ease of purchasing), so two
47uH inductors with a saturation current of at least
1.12A are suitable.
If too much current flows in the inductor, the
ferrite that the inductor is wound on saturates and
the inductor loses its inductive properties. From
the equation
if the inductor value falls, the current ramp
increases causing the ferrite to further saturate…
Therefore must make sure that the inductor never
saturates.
The Wurth 744071470 is a 47uH inductor with a 1.3A
saturation current.
744071470 Datasheet
Rsense Calculation
The current sense resistor calculation for a Cuk
converter is different than that of the other dc/dc
converter topologies. When the MOSFET switches on,
the current flowing in the current sense resistor is
equal to the sum of the two inductor currents (see
Output Diode Choice for a further explanation). The
sense resistor has to be scaled accordingly.
Thus with a peak sense current of 706mA + 1.12A
(=1.83A) and a current sense trip threshold of 120mV
(see LT3757 datasheet), a current sense resistor of
50mOhms should ensure good circuit operation.
On startup, when the MOSFET first turns on, there is
a possibility that the peak current in the input
inductor could ramp to well beyond its saturation
rating. Given that we have a current sense resistor
of 50mOhms and a sense voltage level of 120mV, it is
conceivable that the peak current could reach 2.4A.
However the LT3757 is fitted with a Soft Start
function that gradually increases the peak current
threshold comparator limit, determined by a
capacitor on the SS pin. Appropriate sizing of this
capacitor ensures the circuit’s output voltage has
settled (thus the inductor currents have settled),
before the threshold comparator reaches its full
value.
MOSFET Choice
The MOSFET needs to be able to handle the peak
current in both inductors (1.83A) so in this design
a drain source current rating (Id) of 5A is more
than sufficient. The Drain–Source voltage (Vds)
needs to be in excess of Vcap (= Vin + |Vout|), so
anything above 25V is suitable for a 10V input and
-5V output.
The Gate-Source turn on voltage of the MOSFET (Vgs)
needs to be less than the input voltage, to ensure
that the voltage coming out of the Gate pin can
actually activate the MOSFET. Logic level MOSFETs
have a low turn on voltage, are widely available and
usually perfect for low voltage dc/dc converters.
The above parameters represent the bare minimum
characteristics of the MOSFET. However, to get a
good design, we must ensure that the losses in the
MOSFET are as low as possible. The MOSFET switch
presents 2 losses in the circuit: Switching losses
and conduction losses.
The switching losses result from current flowing
through the MOSFET at the same time that a voltage
is across the MOSFET (so power is generated in the
MOSFET), during the turn on and turn off times of
the MOSFET. For a given gate drive coming out of the
controller IC, the lower the Gate-Source capacitance
of the MOSFET, the quicker the MOSFET will turn on.
Thus the Qg specification of the MOSFET is important
and should be as low as possible. The Qg of the
MOSFET will also have an impact on the heat
dissipation of the chip, especially if the input
voltage to the chip is high.
Charge is dictated by the equation:
Charge (Q) = Current (I) x Time (s)
Since Frequency is the inverse of Time, we can write
So we can calculate the current needed to flow into
the chip, just to charge the gate capacitance of the
FET. Since heat is the product of voltage and
current, if the gate charge is high and/or the
switching frequency is high, the heat dissipation in
the chip will be high.
Once the MOSFET has switched on, the MOSFET presents
a small dc resistance between its Drain and Source
terminals. This is the MOSFETs ‘Drain Source on
resistance’ or Rdson. Again, this needs to be as low
as possible.
Now, MOSFET manufacturers reduce the ON resistance
of the MOSFET by constructing many parallel
conduction paths between the Drain and Source. Thus,
like connecting resistors in parallel, the ON
resistance comes down with more parallel paths.
However, in connecting Drain Source paths in
parallel, a negative effect is that the Gate Source
capacitance (Qg) is also connected in parallel, so a
low ON resistance (and hence low conduction loss)
sometimes implies a high gate source capacitance
(hence high switching loss). Thus the MOSFET that is
chosen should be a compromise between these two
characteristics. In addition, high current MOSFETs
tend to come in much larger packages, so meeting the
ideals of low ON resistance and low Qg might violate
a space requirement spec, so the selection process
has to start over. Engineering, as ever, is a
compromise.
Indeed looking at the selection tables of the MOSFET
manufacturers, it is better to select a MOSFET with
a low ON resistance (less than 10mOhms), then filter
this selection to remove MOSFETs with a Qg of
greater than 10nC then select a MOSFET from this
list, as long as the Gate turn on voltage, Vds and
Id can be met. Starting by selecting MOSFETs with a
Vds of between 20V and 30V might rule out some
excellent devices and compromise the efficiency.
The Fairchild FDS6680 represents a good compromise
between low ON resistance and low gate charge, but
its SO8 package is large and therefore might be
unsuitable for compact designs.
FDS6680 Datasheet
Output Diode Choice
The output diode needs to have the lowest voltage
drop possible to give the lowest power dissipation
(and hence the lowest loss). A Schottky diode is an
ideal choice. During the input inductor charge
phase, the diode is exposed to a reverse voltage of
Vcap, which we have determined is equal to Vin +
|Vout|, thus the reverse breakdown voltage of the
diode should be higher than Vcap.
To calculate the diode current we need to first
consider the current in the output inductor. The
voltage on the anode of the diode oscillates from 0V
(assuming 0V drop across the diode) to –Vcap where –Vcap
is more negative than Vout. To keep a negative
voltage on the output capacitor, the average current
flowing in the output inductor must flow towards the
diode (from right to left through L2 in FIG 11). If
the ripple current in the output inductor is low
compared to the average current, the current in L2
will not fall to zero so there will always be a
current flowing in inductor L2 from Vout towards the
diode.
When the MOSFET switches off, the current from the
input inductor, L1, flows into the diode. In
addition, the current from the output inductor will
also flow through the diode. Therefore the total
diode current during the discharge phase of the
input inductor is equal to the peak current from
both inductors. The diode current rating should be
select accordingly. Our peak current is 1.83A, so
the MBRS340 is a good choice.
MBRS340 Datasheet
Output Capacitor Choice
In continuous conduction mode, the capacitor has a
continual current flowing into it from the output
inductor. Unlike a boost converter, the output
capacitor in a buck regulator does not have to hold
up the output while the inductor is being charged.
The output is made up of 2 components: the ripple
current from the output inductor producing a voltage
across the effective series resistance (ESR) of the
output capacitor and the ripple current charging the
output capacitor according to the equation
Unlike a boost converter where the rectifier diode
current jumps from 0A to the peak inductor current
as the MOSFET switches off, the ripple in a buck
architecture is determined by the ripple
current amplitude, not the peak inductor
current.
Recent innovations in ceramic capacitor design mean
that very low ESR capacitors are available with high
capacitance values. Ceramic capacitors have a
typical ESR of 10mOhms.
Failing that, low ESR tantalum capacitors are
available in much higher capacitance values with ESR
of upwards of 50m Ohms. Of course capacitors can
also be paralleled to increase the capacitance and
reduce the ESR.
In our example the inductor ripple current is 236mA
and the ESR is of a typical tantalum capacitor is
70m Ohms, giving an ESR ripple of 16.5mV.
To calculate the charging ripple, from the equation
above we can see
FIG 12
FIG11, shows the output inductor ripple current (in
red), output voltage ripple (in green) and output
capacitor current (in purple). For convenience the
output capacitor ESR has been reduced to 0 Ohms to
fully illustrate the effect of discharge ripple. It
can be seen that the capacitor current has the same
amplitude as the inductor ripple current, but does
not have the dc offset current (of approx. 1A). This
is easy to picture, since the output current is
equal to the average inductor current (i.e. a
straight line drawn through the middle of the
inductor current) and any current that does not flow
into the load must flow in and out of the capacitor.
To obtain the capacitor current, just subtract the
output current.
Now, we can see that while the capacitor current is
positive (above the dotted white line) the output
capacitor voltage goes up and while it is negative,
the output capacitor voltage goes down. To work out
the amplitude of the ripple voltage on the output
capacitor, we must calculate the average of the
positive part of the capacitor current (above the
dotted white line). Since we know the peak to peak
ripple current (is equal to the inductor ripple
current), the peak ripple current is Iripple/2 and
hence the average of this current (since the current
is triangular) is Iripple/4. We can now work out the
charging ripple.
From
We can see that dt is equal to half the period, so
we can say
Since our capacitor current is positive for half the
ON time and half the OFF time, the above equation
holds true regardless of duty cycle.
Let’s assume we want a ripple voltage of 1% (50mV).
We already have 16.5mV of ripple as a result of the
capacitor ESR, so we now have to have a charging
ripple of 33.5mV
If our ripple current is 236mA and we are operating
at a switching frequency of 300kHz, a capacitor of
3.3uF should suffice. Comparing this to the circuit
in FIG 8, we can immediately see that for the same
output current, the Cuk Converter has much less
output capacitance. This is due to the fact that the
output inductor current continually flows into the
load whereas the output capacitor in the single
inductor inverter has to keep the load current alive
while the inductor is being charged.
Other Points
The feedback resistor values can be calculated using
this spreadsheet:
Feedback Resistor Calculator
The final LTspice circuit is shown in FIG 13 and can
be downloaded here: Cuk
Converter
FIG 13
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Technology Corporation |
